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Dedicated Sensor Conditioner ICs

Dedicated Sensor Conditioner ICs

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Dedicated Sensor Conditioner ICs

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  1. Dedicated Sensor Conditioner ICs Part 1: Op-Amp Instrumentation Amp 4 – 20mA Transmitter Part 2: Temperature Pressure Current Measurements Arthur A. Kay Senior Applications Engineer High Performance Linear Burr-Brown Products Texas Instruments Incorporated 6730 South Tucson Blvd, MS 5005 Tucson, Arizona 85706 (520) 746-6072

  2. Op-Amps

  3. What Causes some of these common concerns? What are the modern solutions? Offset Offset Drift Swing to Rail Noise

  4. Noise

  5. Noise Categories • Extrinsic Noise— • 50 / 60 Hz Line Noise • RFI / EMI • Microphonic • Charge displacement (very hi-Z circuit) • Switching power supplies • Digital noise • Ground loops • Atmospheric • Cosmic • Intrinsic Noise— • Broadband Noise • Thermal Noise • 1/f Noise • Shot Noise • Popcorn (Burst) Noise We’ll concentrate on these types of random noise

  6. Thermal Noise Also known as “Johnson Noise”– first observed by Schottky in 1918, first measured by Johnson in 1928 & soon after formalized by Nyquist. Thermal noise is produced by random motion of charges. The mean- square open- circuit voltage (e) across a resistor (R) is: e2 = 4kTRΔf where k is Temperature (ºK) R is Resistance (Ω) f is frequency (Hz) k is Boltzmann’s constant (1.38E-23 joule/ºK) e is volts (VRMS) e = √4kTRΔf (when Δf = 1Hz, "e" is the voltage noise spectral density)

  7. Thermal Noise vs. Resistance (at 3 Temperatures)

  8. Spectral Noise Density ChartUsed for Op Amp Noise Calculation

  9. Total Noise Equation enT = √[(en1/f)2 + (enBB)2] where: enT =Total rms Voltage Noise in volts rms en1/f = 1/f voltage noise in volts rms enBB = Broadband voltage noise in volts rms

  10. en1/f = (e1/f@1Hz)(√[ln(fH/fL)]) where: en1/f = 1/f voltage noise in volts rms over frequency range of operation e1/f@1Hz = voltage noise density at 1Hz; usually in nV/√Hz fH = upper frequency of frequency range of operation fL = lower frequency of frequency range of operation e1/f@1Hz = (e1/f@f)(√f) where: e1/f@f = voltage noise density at f; usually in nV/√Hz f = frequency of known noise voltage density 1/f Noise Equation

  11. Noise Calculation Continued enT = √[(en1/f)2 + (enBB)2] We’ve learned how to compute this component. How do we compute the broadband noise component?

  12. where: fP = roll-off frequency of pole or poles fBF = equivalent brickwall filter frequency Real Filter Correction vs Brickwall Filter

  13. AC Noise Bandwidth Ratios for a Cascade of n, Low-Pass Filters BWn = (fH)(Kn) Effective Noise Bandwidth Real Filter Correction vs Brickwall Filter

  14. Broadband Noise Equation BWn = (fH)(Kn) where: BWn = noise bandwidth for a given system fH = upper frequency of frequency range of operation Kn = “Brickwall” filter multiplier to include the “skirt” effects of a low pass filter enBB = (eBB)(√[BWn]) where: enBB = Broadband voltage noise in volts rms eBB = Broadband voltage noise density ; usually in nV/√Hz BWn = Noise bandwidth for a given system

  15. Example noise Calculation Given: Input Voltage Noise curve and a single pole system with a roll-off at 3kHz Find: Total rms voltage noise referred to the op amp input

  16. Example Noise Calculation Voltage Noise Calculation: 1/f Voltage Noise Component: e1/f@1Hz = (e1/f@f)(√f) e1/f@1Hz = (20nV/√Hz)(√0.1Hz) = 6.32nV/√Hz en1/f = (e1/f@1Hz)(√[ln(fH/fL)]) en1/f = (6.32nV/√Hz)(√[ln(3kHz/0.1Hz)]) = 20.29nV rms Broadband Voltage Noise Component: BWn ≈ (fH)(Kn) (note Kn = 1.57 for single pole) BWn ≈ (3kHz)(1.57) =4.71kHz enBB = (eBB)(√BWn) enBB = (3nV/√Hz)(√4.71kHz) = 205.9nV rms Total Voltage Noise (referred to the input of the amplifier): enT = √[(en1/f)2 + (enBB)2] enT = √[(20.29nV rms)2 + (205.9nV rms)2] = 206nV rms

  17. Calculating Noise Vpp from Noise Vrms Relation of Peak-to-Peak Value of AC Noise Voltage to rms Value *Common Practice is to use CF=6

  18. Noise Temporal Behavior P(f) = 1 / fbeta Power spectral density is independent of frequency for white noise, beta = 0 1/f noise spectrum produced when beta = 1, equal power per octave

  19. Offset

  20. CMOS vs Bipolar CMOS Advantages Low input bias current Small Package Low Cost Allows for Mixed signal BIPOLAR Advantages Low Offset and Low Offset Drift Low Flicker Noise (1/f) CMOS Disadvantages Higher Offset and Drift Higher Flicker Noise (1/f) BIPOLAR Disadvantages High input bias current Poor Swing to the Rail

  21. Bipolar CMOS 3.3uV/°C per milliVolt of Vos + + Vos 0 Vos 0 _ _ Temperature Temperature Offset and Offset Drift – CMOS vs. Bipolar The offset drift of a CMOS amplifier is not predictable. This makes it difficult to compensate for the error at the manufacturer.

  22. Auto-Zero – The modern solution to offset and offset drift in CMOS amplifiers. Digital circuitry is used to continuously calibrate the device so that the offset is nearly zero. Two “nulling” amplifiers (AN1 and AN2) are used in an alternating two cycle sequence to create an ideal “offset-free” op-amp. One of the nulling amplifiers will sample the offset while the other uses its previously sampled offset to act as an ideal zero offset, low frequency op-amp. A wide band (AF) amplifier is used in parallel with the nulling amplifiers. Its offset is not corrected for, but is minimized because its low frequency gain is substantially lower then the gain of the nulling amplifiers.

  23. Comparing Auto-Zero to other Topologies Auto-Zero CMOS – OPA334 Max Offset = 5uV Max Offset Drift = 0.05uV/oC Typical CMOS – OPA363 Max Offset = 500uV Typical Offset Drift = 3.0uV/oC Very High Performance BIPOLAR – OPA277 Max Offset = 20uV Max Offset Drift = 0.1uV/oC

  24. Auto-Zero noise versus Standard Op-Amp

  25. Swing to the Rail

  26. Traditional Rail to Rail Input Topology OPA703 The traditional approach uses n-channel and p-channel MOSFET’s in parallel to achieve swing beyond the rails. The problem is that there is a transition zone where both pairs of transistors are on. The PSR, CMR, offset voltage, and offset drift are different from normal in this region.

  27. Swing to the Rail – Modern Single Supply Rail-to-Rail – OPA363 The modern approach uses a charge pump to eliminate the need for the parallel P-Channel and N-Channel MOSET’s used in the traditional approach. Thus, the transition zone where offset is disturbed is eliminated.

  28. Rail to Rail Output An Op-Amp with Rail-to-Rail output can swing within a few millivolts of the rail when it is lightly loaded. The output stage of the amplifier prevents the output from reaching the rail exactly. Also for increasing load, the output swing to the rail will degrade. Below are examples showing the effect of load on output swing. OPA703 OPA364

  29. Open-Loop Gain is 1/Slope between end-points (in dB). VOS VOUT Max Vout Min Vout Linear Operation Near Rail AOL = ΔVOUT / ΔVOS

  30. Instrumentation Amps

  31. Why An Instrumentation Amplifier (IA)? IAs Reject Common Mode & Amplify Differential Input Voltages • Why IA ? • Input Signal has: • High Common Mode & Small Differential • Common Mode Noise • IA provides: • High Common Mode Rejection • Accurate Differential Gain (1-10,000) • High Input Impedance - Low Signal Load

  32. IA Technical Concepts • Common Mode Voltage • Don’t get Bit by the “Snake in the Grass” = Old CM • Differential Voltage • Inside View of IA Topologies • 3 Op Amp • 2 Op Amp • 2 Op Amp with Output Gain Amplifier • 3 Op Amp V-I to I-V

  33. The Real “Common Mode” View Some People Say: Common Mode Voltage is +5V Differential Voltage is +3V IA / System View: Common Mode Voltage is +5V Differential Voltage is +3V

  34. The “Inside” View of IAs 3 Op Amp 3 Op Amp with V-I to I-V 2 Op Amp 2 Op Amp + Gout

  35. 3 Op Amp IA Implementation Symmetrical topology yields high CMR Accurate Gains with Single Gain-Set Resistor Gain range from 1 –10,000 Common Mode Voltage Reduced at high gains due to A1, A2 out Swing and VREF

  36. 2 Op Amp IA Implementation Asymmetrical topology yields decreased CMR due to unequal phase shifts from VIN to VOUT Limited accurate Gain Range (10 or 50) for CMOS due to internal absolute resistor tolerances (+/-18%), High Tempco (+/-700ppm/C) Common Mode Voltage reduced by VREF Low Cost CM Range to Negative Rail

  37. Gain Accuracy vs. External Resistor For 2 Op Amp CMOS Implementation Gain = 10 (Gain Pins Open)  Gain Error = 0.1% Gain = 20 (RG= 30K)  Gain Error = 1.3% Gain = 30 (RG= 10K)  Gain Error = 1.2% Gain = 40 (RG= 3.3K)  Gain Error = 0.65% Gain = 50 (Gain Pins Connected)  Gain Error = 0.25%

  38. 2 Op Amp + Gout Asymmetrical topology yields decreased CMR due to unequal phase shifts from VIN to VOUT Gain Accuracy dominated by Accuracy of Two External Resistors Gain range from 5-1000 Common Mode Voltage reduced by VREF Low Cost CM Range to Negative Rail

  39. High CMR Large CMV Range (R-R Input) Gain Accuracy dominated by Two External Set Resistors Wide Gain range from <0.1 to 10000 Output Offset set by resistor divider w/o CMR effects Low Cost Precision –Low Offset, Low Drift, Low1/f Noise 3 Op Amp w/V-I to I-V

  40. 3 Op Amp V-I to I-V – New Technology INA326/INA327; INA337/INA338 Patent Pending Topology Yields: Rail-Rail I/O Single Supply Low Offset, Low Drift, Low 1/f Noise: A1, A2, A3 are “auto zeroed” for low offset, drift, 1/f noise Gain Accuray - Current Mirrors: Gain & Drift & NonLinearity direct result of mirror accuracy Dominant internal error source is current mirror matching New and Unique current mirror topology – patent pending! No trims! Current mirrors are auto-calibrating! Application Info: Gain Accuracy dominated by 2 External Resistors VOUT Offset set by high impedance pin “R2” VOUT=2(R2/R1)VIN

  41. INA326/INA327 & INA337/INA338 INA326 / INA327INA337 / INA338 Offset Voltage+100uV +100uV Offset Drift+ 0.4uV/C + 0.4uV/C Voltage Noise0.8uVp-p typ (0.01Hz-10Hz) 0.8uVp-p typ (0.01Hz-10Hz) Input Voltage Range-0.02V to (VS+ 0.02V) +0.25V to (VS + 0.1V) High CMR 110dB min @ G=100 110dB min @ G=100 Gain Error +0.2% +0.2% Gain Drift +25ppm/C +25ppm/C NonLinearity+0.01% FS +0.01% FS Shutdown Pin INA327 OnlyINA338 Only Specified Range -40°C to +85°C-40°C to +125°C Single Supply +2.7V to +5.5V +2.7V to +5.5V Dual Supply+/-1.35V to +/-2.75V +/-1.35V to +/-2.75V Package8MSOP (INA326) 8MSOP (INA337) 10MSOP (INA327) 10MSOP (INA338) Price @ 1k $1.70 / $1.85$1.71 / $1.85

  42. INA326 Low Noise, No 1/f Noise Super Noise Performance! Note this axis is often shown in uV. Gain = 100, Noise RTI

  43. Current Transmitters

  44. 2-Wire 4-20mA • Power & Signal are on the Same Pair of Wires • Power provide by the Receiver Side • Signal Approach uses Current because it's Immune to Noise • XTR Conditioner “floats” above VOUT (IOUT x RL)

  45. 3-Wire 4-20mA • Power Provide by the XTR (Transmitter) side • Signal Approach uses Current because it's Immune to Noise • XTR Conditioner Referenced to XTR Gnd • Receiver Load Referenced to XTR Gnd

  46. 4-20mA Transmitter Design Solutions • 2-Wire General Purpose • 3-Wire General Purpose • 4-20mA Current Loop Receiver • 2-Wire RTD Conditioners • with Linearization • 2-Wire Bridge Sensor Conditioners • with Linearization • 2-Wire RTD Conditioner • with Digital Cal (Linearization, Span, Offset)

  47. Temperature Sensors

  48. Different Temperature Sensors

  49. Thermistor • Typically made out of semiconductor material • Often have high output resistance and large changes vs. temperature • Very Nonlinear • Temp range limited (typically less then 150C)

  50. Thermistor Application – Control Loop