1 / 46

TWEPP – Vienna – September 27 th 2011

1. 1. Wideband (500 MHz) 16 bit Dynamic Range Current Mode Preamplifier for the CTA cameras. D. Gascón a , A. Sanuy a , J. M. Paredes a , Ll. Garrido a , M. Ribo a , X. Sieiro b Universitat de Barcelona Institut de Ciències del Cosmos ICC-UB (a) Departament d’Electrònica (b).

carnig
Télécharger la présentation

TWEPP – Vienna – September 27 th 2011

An Image/Link below is provided (as is) to download presentation Download Policy: Content on the Website is provided to you AS IS for your information and personal use and may not be sold / licensed / shared on other websites without getting consent from its author. Content is provided to you AS IS for your information and personal use only. Download presentation by click this link. While downloading, if for some reason you are not able to download a presentation, the publisher may have deleted the file from their server. During download, if you can't get a presentation, the file might be deleted by the publisher.

E N D

Presentation Transcript

  1. 1 1 Wideband (500 MHz) 16 bit Dynamic Range Current Mode Preamplifier for the CTA cameras D. Gascóna, A. Sanuya, J. M. Paredesa, Ll. Garridoa, M. Riboa, X. Sieirob Universitat de Barcelona Institut de Ciències del Cosmos ICC-UB (a) Departament d’Electrònica (b) TWEPP – Vienna – September 27th 2011

  2. Introduction PreAmplifier for CTA (PACTA) Input stage Current gain stage Transimpedance stage Noise Test results Conclusions. Outlook 2

  3. I. Introduction: Cherenkov telescopes Gamma ray Air shower ~ 10 km ~ 1o Cherenkov light ~ 120 m Stereoscopy provides better: • Angular resolution. • Energy resolution (height). • Background rejection. • Sensitivity.

  4. I. Introduction: the Cherenkov Telescope Array (CTA) observatory • How CTA aims to extend energy range and increase sensitivity? • Large array (>1 km2)of Cherenkov telescopes (50-100): over 100K channels • Different sizes: dish from 6 to 24 m • Camera and electronics must be optimized in terms of • Performance • Cost and reliability: integration

  5. I. Introduction: the camera Front end electronics: Pixel: fast photosensors High QE PMTs (baseline) // SiPM Modularity: cluster of 7/8 pixels Front end electronics in the camera Digitization & trigger Huge dynamic range: 16 bits Signals up to 5 Kphe Single phe resolution for calibration: Noise  1/8 phe PMT gain 40Ke  ENC  5Ke (10 ns integration) 5 HESS cluster • High BW (>300 MHz, full chain): • Night Sky Background: • Up to 100 MHz • Minimize integration time Simulation (S. Vorobiov) This work

  6. I. Introduction: requirements To develop a generic preamplifier valid for any CTA camera Common component: low cost and reliability for mass production This component must fulfil a set of demanding requirements: 6

  7. II. PreAmplifier for CTA (PACTA): circuit design Basic circuit: Super common base input Low noise / High DR: used in LHC calorimetry Cascode current mirror with CB feedback Fully differential transimpedance amp. (TIA) Performances BW > 500 MHz Low Zi < 10 Ohm up to > 500 MHz Low noise (in=10 pA/sqrt(Hz)) Differential: optimal CMRR and PSRR But the current mirror can not stand a 1000 phe pulse Limited to 12/13 bit DR Saturation at 500 to1000 phe Not enough for 16 bit… 7 7 Simplified schematic

  8. II. PreAmplifier for CTA (PACTA): circuit design Previous circuit is modified to split the input current: Current is divided in the common base stage Different current mirrors for high and low gain Each can be optimized for BW / linearity Dedicated saturation control circuit is added to the HG mirror Current division remains operational even if HG mirror saturates Saturation threshold of HG mirror can be controlled Range: > 6000 phe True delta pulse with 500 MHz BW No arrival time effect considered 8 8 Patent pending Simplified schematic Single ended and differential versions

  9. II. PreAmplifier for CTA (PACTA): prototypes AMS SiGe 0.35 um techno: High speed, low noise and offset HBTs 5 V / 30 GHz NPN HBTs 3.3 V power supply: DR vs power trade off 9 9 PACTAv1.2 chip 2 mm2 QFN32 package Submitted June 2011 PACTAv1.1 chip 2 mm2 QFN32 package Back from foundry Oct. 2010 • 2 single ended PACTAs • 1 differential PACTA • Main modifications • New OpAmp (s.e. and fully diff. versions): • Low output impedance stage • Higher Slew Rate (1 V/ns) • Improve compensation of input stage and current amplifier • An input stage + current amp. block • A complete differential PACTA: • No low impedance driver

  10. III. Input stage Super common base: low Zi (<15 ) Two unbalanced current outputs: Vbe of Q1 and Q0 is the same Q1 = 14 * Q0: current splitting Cascode transistors to improve linearity: Minimize variation of Vc of Q1 and Q0 Minimize Early effect HF feedback loop (Return Ratio T(s)): Miller compensation Cc: pole splitting Q1/Q0 emitter degeneration RE: Linearizes Zi and improves matching Limited by DR and noise: tradeoff 10 10 Phase Margin of T(f) vs vccb GBWP of T(f) vs vccb T(f) (mag) vs Cc Cc=750 fF Cc=0 T(f) (phase) vs Cc With pole splitting: Dom. Pole Zero in Zi(s) At f >2 GHz

  11. III. Input stage Input impedance depends on Vccb gmQf 11 11 • No impact of dominant pole on Zi • Dominant pole also affects open loop Zi • No inductive effect in the BW of interest zZi Using Blackman´s impedance formula: pZi Zi(f) (mag) vs vccb Zi(f) (mag) vs Cc pZi zZi Zi(f) (phase) vs vccb Zi(f) (phase) vs Cc

  12. III. Input stage Bonding inductance must be considered: QFN32 package: < 1mm bond wires Series resonance: CPAD*Lb (input induct.) Rs to increase  (damping factor) On chip decoupling cap. also need damp. res. Ground inductance Lg is critical: Direct feedback to Qf emitter Output driver provides large current pulses: Ringing or oscillation possible! Downbonds to package cavity < 0.5 mm wires Multiple grounds, different grounds for: Input stage TIAs 12 12 T(f) (mag) vs Lb T(f) (phase) vs Lb T(f) (mag) vs CC T(f) (mag) vs Lg Nominal inductances T(f) (phase) vs Lg T(f) (phase) vs CC

  13. IV. Current amplifier: saturation control circuit Low voltage cascode current mirrors Local feedback: common base HBT Qcb Minimize input impedance Minimize Voltage variation of VCQ1 Wideband amplifier with high DR: BW > 500 MHz Current gain (AI): 2.5 > 12 bit DR But M1/M1c to ohmic region for large signals: Feedback and low input impedance are lost Saturation ctrl circuit “quenches” large signals: Qoc1 is controlled by the voltage Vc-Vm For low currents Vc-Vm<<Vbe_On If drain current of M1 increases, Vc increases and Vm decreases Turn on point of Qoc1 can be tuned through Vlim 13 13

  14. IV. Current amplifier: saturation control circuit Feeback loop must be compensated Two main poles related to M1 gate and input nodes Add Cc for dominant pole comp Drawback: limits amplifier BW BW limited by Cin Must increase Cc to compensate BW > 500 MHz provided that: Cin < 700 fF With PM > 65 deg Source degeneration (RSM) in v1.2 For large bias currents (or with a high pulse rate) gmM1 increases GBW increases and PM decreases Source degeneration limits the effective value of gmM1 14 14

  15. V. TIA Single ended and fully differential versions OpAmp architecture: HBT input pair + folded cascode Miller gain stage Output stage: None for PACTAv1.1 Low impedance class AB (v1.2) Differential OpAmp with CMFB: Gain > 65 dB GBW> 700 MHz PM > 70 deg SR = 1 V/ns (PACTAv1.2) Low output impedance push-pull stage: Based on NPN HBTs Feedback loop must be extremely fast Compensation is critical: Operates in closed loop (OpAmp output stage) Drives 50 Ohm loads (AC coupled) 15 15 Thanks to J. Lecoq, E. Delagnes and P. Moreira Class AB output stage Provides up to 20 mA peak current with 5 mA quiescent current Compensation of the local FB loop Return Ratio T(f) versus Vbef PM of T(f) UnityGainFreq of T(f) GBW of T(f) Post layout simulation including bonding inductances GM of T(f)

  16. VI. Noise For the single ended version Series Noise Dominated by the input stage Paralllel Noise: Significant contribution of Input stage Current mirrors Parallel noise dominates for Cin<5 pF 16 16

  17. VI. Noise Differential version: uncorrelated noise sources add in quadrature Noise should be sqrt(2) higher Integrated output noise referred to an input current noise source Equivalent noise current (ENI) Minimal for low cap. sensors (PMT): in Still quite ok for high capacitance sensors 17 17 • en=1.13 nV/sqrt(Hz) SiPM, LAAPD PMT • in=11.8 pA/sqrt(Hz)

  18. VII. PACTAv1.1 chip test results Pulse shape 18

  19. VII. PACTAv1.1 chip test results Pulse shape (normalized waveforms) 19

  20. VII. PACTAv1.1 chip test results Frequency response Small signal BW exceeds 500 MHz both for high gain and low gain Minimize peaking for LG in final version 20

  21. VII. PACTAv1.1 chip test results Transimpedance gain (integral of the pulse) and linearity HG about 1 KOhm LG about 50 Ohm Relative non-linearity error < 2 % 100x(Meas-Fit)/Fit 21 100 phe 1 phe

  22. VII. PACTAv1.1 chip test results Dedicated board with PMT (R81619mod) Additional voltage gain (14.4 V/V) Total gain TIA gain is 14.4 KOhm To minimize the impact of following stages on the input ref. noise Differential to single ended conversion + 50 Ohm driver Lot of work on grounding and shielding Many thanks to: R. Mirzoyan, D. Fink, P. Nayman and F. Toussenel 22

  23. VII. PACTAv1.1 chip test results Single photoelectron spectra at 900 V PMT at 900 V, gain about 4x104 New analysis method (R. Mirzoyan) with less systematic under development Integration time: 10 ns ENC of about 5000 electrons (11.8 pVs / q x 14.4 K) S/N = 8 23 PACTA + NECTAr0 chip (see poster session) PACTA + 20GS/s DPO Scope

  24. VII. PACTAv1.1 chip test results Input referred noise as function of the integration time Theoretical and measured noise (with and without PMT) Still some few hundred e contribution of pick up noise Differential configuration to minimize CM noise With single ended configuration theoretical ENC should be 1/sqrt(2) 24

  25. VII. PACTAv1.1 chip test results Preliminary tests with SiPM Low Zin current mode circuit are well suited for SiPM readout DC coupling without external components We just took an available MPPC (S10931-050P) 1 V overvoltage Recovery time seems to be dominated by internal SiPM time constant 25 20 ns

  26. VIII. Conclusions Wideband and high dynamic range current mode preamplifier for CTA cameras PACTAv1.1 meets most of the requirements : Input Referred Noise< 400 nA rms SNR for SPE spectra: 8, at the nominal PM gain (40K) Input range: > 20 mA peak Dynamic range: 15.9 bits Relative linearity error for charge measurements: < 2 % BW: 500 MHz. Both for HG and LG Input impedance: 10 to 15 Ω. For full BW. A cable / tline driver has been implemented in PACTAv1.2 Single ended and differential versions Power consumption: 150 mW Factor 3 or 4 smaller than current prototypes build with COTS New versions of the circuit for SiPM readout are under development Potentially, good time resolution 26 26 26 26 TWEPP – Vienna – September 27th 2011 D.Gascón

  27. Back-up 27

  28. 28

  29. 29

  30. V. Noise: time domain analysis 30 • The noise variance is: • For T>>: Approximation error as function of 

  31. II. Noise requirements Good single photoelectron resolution: S/N > 10 in the charge spectra How to translate to typical specification for amplifiers? Series (en) and parallel (in) noise power spectral densities Assumptions Flicker noise is negligible Current preamp: RPM open Voltage preamp: ZT resistive (RPM) CPAR is small enough: series noise negligible for I amp RPM is small enough: parallel noise negligible for V amp Noise variance at the output of gated integrator is (approx): T is the integration time 31 ZT = RPM // CPAR V preamp: I preamp:

  32. II. Noise requirements For a voltage preamplifier, the signal at the output of gated integrator is Zi is the PM load impedance And the S/N is Max input referred noise of the amplifier (en): For a S/N > 10 32 Voltage amplifier Current amplifier < 10 pA/sqrt(Hz) ! < 1 nV/sqrt(Hz) !

  33. III. The ATLAS LAr preamplifier Low noise Series noise: en=0.36 nV/sqrt(Hz) Parallel noise: in=6.7 pA/sqrt(Hz) Low gain option: 40 K Low noise preamplifier is needed High dynamic range: about 14 bits Super-common base: Small input impedance Photo-detector current is sensed 33

  34. IV. Measurements Single photoelectron spectra Pulser LED @ 460 nm < 500 ps FWHM Afterpulsing observed LASER @ 640 nm < 50 ps FWHM Optical attenuator Hamamatsu R7600 PM To be repeated with R9420 34

  35. IV. Measurements Single photo-electron spectra @HV = 500 V : 40 K gain Preamp noise:3-4 Ke, limited by common mode and pick-up noise 35 PM HV 500 V Single photoelectron

  36. IV. Measurements Effect of the integration time 36

  37. IV. Measurements: conclusions and plans Super common base architecture is promising Low noise High dynamic range Low input impedance However covering full dynamic range with a single preamplifier is still difficult: Preamp saturation (for LAr preamplifier) 5 mA Max signal (6 Kphe) 20 mA peak current @ 40 K gain 37 37

  38. VII. PACTAv1.1 chip test results Low inductance QFN socket Test PCB with socket for characterization Noise/single phe measurement: Dedicated PCB with Additional gain 50 ohm drivers Test set-up: Agilent 81155A pulse generator HP RF signal generator (DC-1GHz) Picoquant Laser Pulser (50 ps FWHM) Tektronix TDS7154B scope: 1.7 GHz 20 GS/s Active differential probe: 1 GHz 38

  39. III. PACTAv1.1 chip test results Transimpedance gain (amplitude) and linearity HG about 1 KOhm LG about 50 Ohm Relative non-linearity error < 3 % 100x(Meas-Fit)/Fit 39 1 phe 100 phe

  40. III. PACTA chip test results PMT signal shape 40 PMT directly to scope PACTA Output HV=1200 V 2.1 ns PACTA Output HV=900 V

  41. III. PACTA chip test results Gain is calibrated for single photoelectron spectra, comparing: Single photoelectron signals with no preamp, direct to scope (only high gain) Single photoelectron with PACTA Result is quite close to what we expect: 14.3 Kohm 41

  42. III. PACTA chip test results S/N ratio for single photoelectron measurements Optimal integration time is the which maximizes S/N: about 10 ns 42

  43. III. PACTAv1.1 chip test results Preliminary tests with SiPM Bi-gain is also working 43

  44. III. PACTA chip test results Preliminary tests with SiPM Charge spectrum 44

  45. PACTA Connections Different scenarios: location and connection to FE PMT cluster as single board No pb to place preamp very close to PMT (< 3 cm) PACTA to FE connection: Differential impedance controlled PCB traces Cheap and robust PMT cluster in several boards: PMT – cable - PACTA PACTA Zin is 10-20 Ohm: no adapted If distance between PMT and PACTA increases: EMC… Preliminary test shows than 10 cm could be ok What is the min length that cluster design can achieve? PACTA to FE connection: Differential impedance controlled PCB traces PMT cluster in several boards: PMT – PACTA – cable(s) PACTA can be very close to PMT but… Room for PACTA + test pulse + monitoring … ??? + CABLES and connectors ??? How many ? NOT RECOMENDED 45

  46. IV. PACTAv1.2 A fully differential TIA and 2 single ended TIAs Each of them with HG and LG outputs New 50 Ohm drivers have been integrated Many thanks to J. Lecoq(LPC), E. Delagnes (CEA/Saclay) and P. Moreira (CERN) Class AB push-pull follower with fast local feedback Dynamic range to directly match std differential ADCs: 1 V for single ended version 2 Vpp for fully differential version Total power consumption : 120 – 150 mW For 2 (High and Low gain) differential outputs To compare with COTS solution Preamp FE amplifiers: double gain + level adaptation Power consumption reduction: 500 mW /ch (aprox) Single ended input stage for minimal noise According to simulations 1/1.2 wrt fully differential TIA Ideally should be 1/sqrt(2) Common mode noise (bias current sources, etc) Minor changes on the current mode input stage: Minimize high frequency peaking Closer to a first order system response 46 46 PACTA1.2 chip SiGe BiCMOS 0.35um AMS 2 mm2 QFN32 package Submitted on June 6th

More Related